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MIC2182
Micrel
MIC2182
High-Efficiency Synchronous Buck Controller Final Information
General Description
Micrel's MIC2182 is a synchronous buck (step-down) switching regulator controller. An all N-channel synchronous architecture and powerful output drivers allow up to a 20A output current capabilty. The PWM and skip-mode control scheme allows efficiency to exceed 95% over a wide range of load current, making it ideal for battery powered applications, as well as high current distributed power supplies. The MIC2182 operates from a 4.5V to 32V input and can operate with a maximum duty cycle of 86% for use in lowdropout conditions. It also features a shutdown mode that reduces quiescent current to 0.1A. The MIC2182 achieves high efficiency over a wide output current range by automatically switching between PWM and skip mode. Skip-mode operation enables the converter to maintain high efficiency at light loads by turning off circuitry pertaining to PWM operation, reducing the no-load supply current from 1.6mA to 600A. The operating mode is internally selected according to the output load conditions. Skip mode can be defeated by pulling the PWM pin low which reduces noise and RF interference. The MIC2182 is available in a 16-pin SOP (small-outline package) and SSOP (shrink small-outline package) with an operating range from -40C to +85C.
Features
* * * * * * * * * * * * * * * * * * * * * * * * * 4.5V to 32V Input voltage range 1.25V to 6V Output voltage range 95% efficiency 300kHz oscillator frequency Current sense blanking 5 impedance MOSFET Drivers Drives N-channel MOSFETs 600A typical quiescent current (skip-mode) Logic controlled micropower shutdown (IQ < 0.1A) Current-mode control Cycle-by-cycle current limiting Built-in undervoltage protection Adjustable undervoltage lockout Easily synchronizable Precision 1.245V reference output 0.6% total regulation 16-pin SOP and SSOP packages Frequency foldback overcurrent protection Sustained short-circuit protection at any input voltage 20A output current capability DC power distribution systems Notebook and subnotebook computers PDAs and mobile communicators Wireless modems Battery-operated equipment
Applications
Typical Application
VIN 4.5V to 30V* R7 100k D2 MIC2182-3.3BSM 10 11 SD103BWS VIN VDD C5 0.1F
6 2
BST EN/UVLO HSD PWM VSW LSD
14 16 15 13 12 8 9 7
C6 0.1F
C9 4.7F 16V
C11 22uf 35V x2 Q2* Si4884 Q1* Si4884 L1 10H D1 B140 VOUT 3.3V/4A C7 220uf 10V x2 GND
R2 0.02
C4 1nF
1 3
SS COMP SYNC
PGND CSH VOUT VREF SGND
4
C3 0.1F C2 2.2nF GND
R1 2k
5
C13, 1nF
* 30V maximum input voltage limit is due to standard 30V MOSFET selection.
C1 0.1F
See "Application Information" section for 5V to 3.3V/10A and other circuits.
4.5V-30V* to 3.3V/4A Converter
Micrel, Inc. * 1849 Fortune Drive * San Jose, CA 95131 * USA * tel + 1 (408) 944-0800 * fax + 1 (408) 944-0970 * http://www.micrel.com
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MIC2182
MIC2182
Micrel
Ordering Information
Part Number MIC2182BM MIC2182-3.3BM MIC2182-5.0BM MIC2182BSM MIC2182-3.3BSM MIC2182-5.0BSM Voltage Adjustable 3.3V 5.0V Adjustable 3.3V 5.0V Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C Package 16-pin narrow SOP 16-pin narrow SOP 16-pin narrow SOP 16-pin narrow SSOP 16-pin narrow SSOP 16-pin narrow SSOP
Pin Configuration
MIC2182
SS 1 PWM 2 COMP 3 SGND 4 SYNC 5 EN/UVLO 6 FB 7 CSH 8 16 HSD 15 VSW 14 BST 13 LSD 12 PGND 11 VDD 10 VIN 9 VOUT SS 1 PWM 2 COMP 3 SGND 4 SYNC 5 EN/UVLO 6 VREF 7 CSH 8
MIC2182-x.x
16 HSD 15 VSW 14 BST 13 LSD 12 PGND 11 VDD 10 VIN 9 VOUT
Adjustable 16-pin SOP (M) 16-Pin SSOP (SM)
Fixed 16-pin SOP (M) 16-Pin SSOP (SM)
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Pin Description
Pin Number 1 Pin Name SS Pin Function Soft-Start (External Component): Connect external capacitor to ground to reduce inrush current by delaying and slowing the output voltage rise time. Rise time is controlled by an internal 5A current source that charges an external capacitor to VDD. PWM/Skip-Mode Select (Input): Low sets PWM-mode operation. 1nF capacitor to ground sets automatic PWM/skip-mode selection. Compensation (Output): Internal error amplifier output. Connect to capacitor or series RC network to compensate the regulator control loop. Small Signal Ground (Return): Route separately from other ground traces to the (-) terminal of COUT. Frequency Synchronization (Input): Optional. Connect to external clock signal to synchronize the oscillator. Leading edge of signal above the threshold terminates the switching cycle. Connect to SGND if unused. Enable/Undervoltage Lockout (Input): Low-level signal powers down the controller. Input below the 2.5V threshold disables switching and functions as an accurate undervoltage lockout (UVLO). Input below the threshold forces complete micropower (< 0.1A) shutdown. Reference Voltage (Output): 1.245V output. Requires 0.1f capacitor to ground. Feedback (Input): Regulates FB pin to 1.245V. See "Application Information" for resistor divider calculations. Current-Sense High (Input): Current-limit comparator noninverting input. A built-in offset of 100mV between CSH and VOUT pins in conjunction with the current-sense resistor set the current-limit threshold level. This is also the positive input to the current sense amplifier. Current-Sense Low (Input): Output voltage feedback input and inverting input for the current limit comparator and the current sense amplifier. [Battery] Unregulated Input (Input): +4.5V to +32V supply input. 5V Internal Linear-Regulator (Output): VDD is the external MOSFET gate drive supply voltage and an internal supply bus for the IC. Bypass to SGND with 4.7F. VDD can supply up to 5mA for external loads. MOSFET Driver Power Ground (Return): Connects to source of synchronous MOSFET and the (-) terminal of CIN Low-Side Drive (Output): High-current driver output for external synchronous MOSFET. Voltage swing is between ground and VDD. Boost (Input): Provides drive voltage for the high-side MOSFET driver. The drive voltage is higher than the input voltage by VDD minus a diode drop. Switch (Return): High side MOSFET driver return. High-Side Drive (Output): High-current driver output for high-side MOSFET. This node voltage swing is between ground and VIN + 5V - Vdiode drop.
2 3 4 5
PWM COMP SGND SYNC
6
EN/UVLO
7 (fixed) 7 (adj) 8
VREF FB CSH
9 10 11
VOUT VIN VDD
12 13 14 15 16
PGND LSD BST VSW HSD
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Absolute Maximum Ratings (Note 1)
Analog Supply Voltage (VIN) ....................................... +34V Digital Supply Voltage (VDD) ......................................... +7V Driver Supply Voltage (BST) .................................... VIN +7V Sense Voltage (VOUT, CSH) ............................. 7V to -0.3V Sync Pin Voltage (VSYNC) ................................ 7V to -0.3V Enable Pin Voltage (VEN/UVLO) ...................................... VIN Power Dissipation (PD) SOP ................................................ 400mW @ TA= 85C SSOP ............................................. 270mW @ TA= 85C Ambient Storage Temperature (TS) ......... -65C to +150C ESD, Note 3
Operating Ratings (Note 2)
Analog Supply Voltage (VIN) ........................ +4.5V to +32V Ambient Temperature (TA) ......................... -40C to +85C Junction Temperature (TJ) ....................... -40C to +125C Package Thermal Resistance SOP (JA) .......................................................... 100C/W SSOP (JA) ........................................................ 150C/W
Electrical Characteristics
VIN = 15V; SS = open; VPWM = 0V; VSHDN = 5V; ILOAD = 0.1A; TA = 25C, bold values indicate -40C TA +85C; Note 4; unless noted Parameter MIC2182 [Adjustable], (Note 5) Feedback Voltage Reference Feedback Voltage Reference Feedback Voltage Reference Feedback Bias Current Output Voltage Range Output Voltage Line Regulation Output Voltage Load Regulation Output Voltage Total Regulation MIC2182-3.3 Output Voltage Output Voltage Output Voltage Output Voltage Line Regulation Output Voltage Load Regulation Output Voltage Total Regulation MIC2182-5.0 Output Voltage Output Voltage Output Voltage Output Voltage Line Regulation Output Voltage Load Regulation Output Voltage Total Regulation Input and VDD Supply PWM Mode Skip Mode Shutdown Quiescent Current Digital Supply Voltage (VDD) Undervoltage Lockout VPWM = 0V, excluding external MOSFET gate drive current IL = 0mA, VPWM floating (1nF capacitor to ground) VEN/UVLO = 0V IL = 0mA to 5mA VDD upper threshold (turn on threshold) VDD lower threshold (turn off threshold) 4.7 4.2 4.1 1.6 600 0.1 2.5 1500 5 5.3 mA A A V V V 6.5V < VIN < 32V, 0 < VCSH - VOUT < 75mV VIN = 6.5V to 32V, VCSH - VOUT = 50mV 25mV < (VCSH - VOUT) < 75mV (PWM mode only) 0mV < (VCSH - VOUT) < 75mV (full load range) 6.5V < VIN < 32V 4.95 4.90 4.85 5.0 5.0 5.0 0.03 0.5 0.8 5.05 5.10 5.150 V V V %/V % % 4.5V < VIN < 32V, 0 < VCSH - VOUT < 75mV VIN = 4.5V to 32V, VCSH - VOUT = 50mV 25mV < (VCSH - VOUT) < 75mV (PWM mode only) 0mV < (VCSH - VOUT) < 75mV (full load range) 4.5V < VIN < 32V 3.267 3.234 3.201 3.3 3.3 3.3 0.03 0.5 0.8 3.333 3.366 3.399 V V V %/V % % VIN = 4.5V to 32V, VCSH - VOUT = 50mV 25mV < (VCSH - VOUT) < 75mV (PWM mode only) 0mV < (VCSH - VOUT) < 75mV (full load range) 4.5V < VIN < 32V 1.25 0.03 0.5 0.6 4.5V < VIN < 32V, 0 < VCSH - VOUT < 75mV 1.233 1.220 1.208 1.245 1.245 1.245 10 6 1.257 1.270 1.282 V V V nA V %/V % % Condition Min Typ Max Units
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MIC2182
Parameter Condition Min Typ Max
Micrel
Units
Reference Output (Fixed Versions Only) Reference Voltage Reference Line Regulation Reference Load Regulation Enable/UVLO Enable Input Threshold UVLO Threshold Enable Input Current Soft Start Soft-Start Current Current Limit Current-Limit Threshold Voltage Error Amplifier Error Sense Amplifier Gain Current Amp Current Sense Amplifier Gain Oscillator Section Oscillator Frequency Maximum Duty Cycle Minimum On-Time SYNC Threshold Level SYNC Input Current SYNC Minimum Pulse Width SYNC Capture Range Frequency Foldback Threshold Foldback Frequency Gate Drivers Rise/Fall Time Output Driver Impedance Driver Nonoverlap Time PWM Input PWM Input Current
Note 1. Note 2. Note 3. Note 4. Note 5. Note 6.
1.220 6V < VIN < 32V 0A < IREF < 100A 0.6 2.2 VEN/UVLO = 5V VSS = 0V VCSH = VOUT -3.5
1.245 1 2
1.270
V mV mV
1.1 2.5 0.1
1.6 2.8 5
V V A A
-5
-6.5
75
100
135
mV
20
2.0
270
300 86
330
kHz %
VOUT = VOUT(nominal) + 200mV 0.7 VSYNC = 5V 200 Note 6 measured at VOUT pin 330 0.75
140 1.3 0.1
250 1.9 5
ns V A ns kHz
0.95 60
1.15
V kHz
CL = 3000pF source sink
60 5 3.5 80 8.5 6
ns ns A
VPWM = 0V
-10
Exceeding the absolute maximum rating may damage the device. The device is not guaranteed to function outside its operating rating. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 25C limits are 100% production tested. Limits over the operating temperature range are guaranteed by design and are not production tested. VIN > 1.3 x VOUT (for the feedback voltage reference and output voltage line and total regulation). See applications information for limitations on the maximum operating frequency.
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MIC2182
MIC2182
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Typical Characteristics
Quiescent Current vs. Temperature
2.0 1.8 CURRENT (mA) 1.6 1.4 1.2 1.0 0.8 0.6 0.4 Skip
Quiescent Current vs. Temperature
1.50
4.0 3.5
CURRENT (mA)
Quiescent Current vs. Supply Voltage
PWM
CURRENT (mA)
1.00 0.50 0 UVLO Mode (mA)
3.0 2.5 2.0 1.5 1.0 0.5 0 0 4 8 12 16 20 24 28 32 INPUT VOLTAGE (V) Skip PWM
-0.50 0.20 0.15 0.10 0.05 SHUTDOWN (A)
0.2 0 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
0 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
Quiescent Current vs. Supply Voltage
1.5
REFERENCE VOLTAGE (V)
VREF (Fixed Versions)
REFERENCE VOLTAGE (V)
VREF (Fixed Versions)
1.260 1.250 1.240 1.230 1.220 1.210 1.200 0 200 400 600 800 1000 LOAD CURRENT (A)
1.0 CURRENT (A) 0.5 0 0.5 -0.5 0.4 0.3 0.2 0.1 0 0 4 8 12 16 20 24 28 32 SUPPLY VOLTAGE (V) UVLO Mode (mA)
1.256 1.254 1.252 1.250 1.248 1.246 1.244 1.242 1.240 1.238 1.236
Line Regulation
Load Regulation
SHUTDOWN (A)
0
4
8 12 16 20 24 28 32 SUPPLY VOLTAGE (V)
VREF (Fixed Versions)
VDD REGULATOR VOLTAGE (V) 1.260
VDD
VDD REGULATOR VOLTAGE (V) 5.0 4.8 4.6 4.4 4.2 4.0
vs. Temperature
Line Regulation
5.00
VDD Load Regulation
REFERENCE VOLTAGE (V)
1.255
4.95
1.250
4.90
1.245
4.85
1.240 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
0
4
8 12 16 20 24 28 32 SUPPLY VOLTAGE (V)
4.80
0
5 10 15 20 LOAD CURRENT (mA)
25
VDD REGULATOR VOLTAGE (V)
4.98 4.96 4.94 4.92 4.90 4.88 4.86 4.84
VDD vs. Temperature
FREQUENCY VARIATION (%)
Oscillator Frequency vs. Temperature
FREQUENCY VARIATION (%) 10 8 6 4 2 0 -2 -4 -6 -8 -10 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C) 1.0 0.8 0.6 0.4 0.2 0 -0.2 -0.4 -0.6 -0.8 -1.0 0
Oscillator Frequency vs. Supply Voltage
4.82 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
4
8 12 16 20 24 28 32 SUPPLY VOLTAGE (V)
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Soft-Start Current vs. Temperature
OVERCURRENT THRESHOLD (V)
Overcurrent Threshold vs. Temperature
0.12
OUTPUT VOLTAGE (V)
Current-Limit Foldback
5 4 3 2 1 0 0 VIN = 5V VOUT = 3.3V RCS = 15m
5.0 4.8 CURRENT (A) 4.6 4.4 4.2 4.0 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
0.11
0.10
0.09
0.08 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
1234567 OUTPUT CURRENT (A)
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MIC2182
MIC2182
Micrel
Block Diagrams
VIN
CIN EN/UVLO
6
VDD
VDD
11
Reference V IN
VBG 1.245V
VIN
10
4.7F D2
SS
1
VBST
14
Control Logic PWM
2
HSD
16
Q2
CBST L1 RCS VOUT COUT
VSW
15
LSD
13
D1 Q1
PGND Current Limit
12
PWM Mode to Skip Mode
0.024V
Skip-Mode Current Limit
0.07V
Low Comp
-2%VBG
PWM OUTPUT
Hysteresis Comp Current Sense Amp
CORRECTIVE RAMP
PWM
RESET
CSH
8
VOUT
VBG
AV = 2
9
SYNC
5
Oscillator
R1 Error Amp FB
7
COMP CCOMP RCOMP MIC2182 [adj.]
3
100k
SGND Gm = 0.2x10-3
4
R2 R1 VOUT = 1.245V 1 + R2 VOUT(max) = 6.0V
Figure 2a. Adjustable Output Voltage Version
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VIN
CIN EN/UVLO
6
VDD
VDD
11
Reference V IN
VBG 1.245V
VIN
10
4.7F D2
SS
1
VBST
14
Control Logic PWM
2
HSD
16
Q2
CBST L1 RCS VOUT COUT
VSW
15
LSD
13
D1 Q1
PGND Current Limit
12
PWM Mode to Skip Mode
0.024V
Skip-Mode Current Limit
0.07V
Low Comp
-2%VBG
PWM OUTPUT
Hysteresis Comp Current Sense Amp
CORRECTIVE RAMP
PWM
RESET
CSH
8
VOUT
VBG
AV = 2 R1*
9
SYNC
5
Oscillator
* 82.5k for 3.3V Output 150k for 5V Output SGND
4
Error Amp
COMP CCOMP RCOMP MIC2182-x.x
3
R2 50k 100k Gm = 0.2x10
-3
VREF
7
Figure 2b. Fixed Output Voltage Versions
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MIC2182
MIC2182
Micrel
Control Loop PWM and Skip Modes of Operation The MIC2182 operates in PWM (pulse-width-modulation) mode at heavier output load conditions. At lighter load conditions, the controller can be configured to automatically switch to a pulse-skipping mode to improve efficiency. The potential disadvantage of skip mode is the variable switching frequency that accompanies this mode of operation. The occurrence of switching pulses depends on component values as well as line and load conditions. There is an external sync function that is disabled in skip mode. In PWM mode, the synchronous buck converter forces continuous current to flow in the inductor. In skip mode, current through the inductor can settle to zero, causing voltage ringing across the inductor. Pulling the PWM pin (pin 2) low will force the controller to operate in PWM mode for all load conditions, which will improve cross regulation of transformer coupled, multiple output configurations. PWM Control Loop The MIC2182 uses current-mode control to regulate the output voltage. This method senses the output voltage (outer loop) and the inductor current (inner loop). It uses inductor current and output voltage to determine the duty cycle of the buck converter. Sampling the inductor current removes the inductor from the control loop, which simplifies compensation.
Functional Description
See "Applications Information" following this section for component selection information and Figure 14 and Tables 1 through 5 for predesigned circuits. The MIC2182 is a BiCMOS, switched-mode, synchronous step-down (buck) converter controller. Current-mode control is used to achieve superior transient line and load regulation. An internal corrective ramp provides slope compensation for stable operation above a 50% duty cycle. The controller is optimized for high-efficiency, high-performance dc-dc converter applications. The MIC2182 block diagrams are shown in Figure 2a and Figure 2b. The MIC2182 controller is divided into 6 functions. * Control loop - PWM operation - Skip-mode operation * Current limit * * * * Reference, enable and UVLO MOSFET gate drive Oscillator and sync Soft start
VIN
CIN
VDD
VDD
11
Reference V IN
VBG 1.245V
CONTROL LOGIC AND PULSE-WIDTH MODULATOR
VIN
10
4.7F D2
VBST
14
HSD
16
Q2
CBST L1 RCS VOUT COUT
VSW
15
LSD
13
D1 Q1
PWM Mode to Skip Mode Q R S
LOW FORCES SKIP MODE 0.024V
PGND
12
PWM COMPARATOR
Current Sense Amp
CSH
8
VOUT
VBG
RESET
CORRECTIVE RAMP
AV = 2
9
R1
Oscillator
COMP CCOMP RCOMP MIC2182 [adj.] PWM Mode
3
Error Amp
FB
7
100k
Gm = 0.2x10-3
R1 VOUT = 1.245V 1 + R2
R2
Figure 3. PWM Operation MIC2182 10 June 2000
MIC2182
A block diagram of the MIC2182 PWM current-mode control loop is shown in Figure 3 and the PWM mode voltage and current waveforms are shown in figure 5A. The inductor current is sensed by measuring the voltage across the resistor, RCS. A ramp is added to the amplified current-sense signal to provide slope compensation, which is required to prevent unstable operation at duty cycles greater than 50%. A transconductance amplifier is used for the error amplifier, which compares an attenuated sample of the output voltage with a reference voltage. The output of the error amplifier is the COMP (compensation) pin, which is compared to the current-sense waveform in the PWM block. When the current signal becomes greater than the error signal, the comparator turns off the high-side drive. The COMP pin (pin 3) provides access to the output of the error amplifier and allows the use of external components to stabilize the voltage loop.
Skip-Mode Control Loop
Micrel
This control method is used to improve efficiency at light output loads. At light output currents, the power drawn by the MIC2182 is equal to the input voltage times the IC supply current (IQ). At light output currents, the power dissipated by the IC can be a significant portion of the total output power, which lowers the efficiency of the power supply. The MIC2182 draws less supply current in skip mode by disabling portions of the control and drive circuitry when the IC in not switching. The disadvantage of this method is greater output voltage ripple and variable switching frequency. A block diagram of the MIC2182 skip mode is shown in Figure 4. Skip mode voltage and current waveforms are shown in figure 5B.
VIN
CIN
VDD
VDD
11
Reference V IN
VBG 1.245V
CONTROL LOGIC AND SKIP-MODE LOGIC
VIN
10
4.7F D2
VBST
14
HSD
16
Q2
CBST L1 RCS VOUT COUT
LOW-SIDE DRIVER ONE SHOT
VSW
15
LSD
13
Q1
PGND
12
Q R S Skip-Mode Current Limit
0.07V
Low Comp
ONE SHOT -2%VBG LOW FORCES PWM MODE
Hysteresis Comp 1%
VBG
Current Sense Amp
CSH
8
VOUT AV = 2
9
R1 FB
7
MIC2182 [adj.] Skip Mode
R1 VOUT = 1.245V 1 + R2
R2
Figure 4. Skip-Mode Operation June 2000 11 MIC2182
MIC2182
VIN VSW 0V IL1 ILOAD 0A Reset Pulse VDD 0V VIN + VDD VHSD 0V VLSD VDD 0V
Micrel
Figure 5a. PWM-Mode Timing
VDD VHSD 0V VDD VLSD 0V VIN VOUT 0V ILIM(skip) IL1 0A Vone-shot VDD 0V VOUT
+1% VNOMINAL -1%
VSW
0V IOUT
0A
Figure 5b. Skip-Mode Timing A hysteretic comparator is used in place of the PWM error amplifier and a current-limit comparator senses the inductor current. A one-shot starts the switching cycle by momentarily turning on the low side MOSFET to insure the high side drive boost capacitor, Cbst, is fully charged. The high side MOSFET is turned on and current ramps up in the inductor, L1. The high side drive is turned off when either the peak voltage on the input of the current-sense comparator exceeds the threshold, typically 35mV, or the output voltage rises above the hysteretic threshold of the output voltage comparator. Once the high side MOSFET is turned off, the load current discharges the output capacitor, causing VOUT to fall. The cycle repeats when VOUT falls below the lower threshold, - 1%. The maximum peak inductor current depends on the skipmode current-limit threshold and the value of the currentsense resistor, RCS.
Iinductor(peak) = 35mV R sense
Figure 6 shows the improvement in efficiency that skip mode makes when at lower output currents.
100 PWM 80
EFFICIENCY (%)
Skip 60 40 20 0 0.01
0.1 1 10 OUTPUT CURRENT (A)
100
Figure 6. Efficiency
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MIC2182 Switching from PWM to Skip Mode The current sense amplifier in Figure 3 monitors the average voltage across the current-sense resistor. The controller will switch from PWM to skip mode when the average voltage across the current-sense resistor drops below approximately 12mV. This is shown in Figure 7b. The average output current at this transition level for is calculated below.
0.012 IOUT(skipmode) = RCS
Micrel
rent-limit threshold is 100mV+35mV -25mV. The currentsense resistor must be sized using the minimum current-limit threshold. The external components must be designed to withstand the maximum current limit. The current-sense resistor value is calculated by the equation below:
RCS =
75mV IOUT(max)
The maximum output current is:
IOUT(max) = 135mV RCS
where: 0.012 = threshold voltage of the internal comparator RCS = current-sense resistor value Switching from Skip to PWM Mode The frequency of occurrence of the skip-mode current pulses increase as the output current increases until the hysteretic duty cycle reaches 100% (continuous pulses). Increasing the current past this point will cause the output voltage will drop. The low limit comparator senses the output voltage when it drops below 2% of the set output and automatically switches the converter to PWM mode. The inductor current in skip mode is a triangular wave shape a minimum value of 0 and a maximum value of 35mV/RCS (see Figure 7b). The maximum average output current in skip mode is the average value of the inductor waveform:
IOUT(max skipmode) = 0.5 x 35mV RCS
The capacitor on the PWM pin (pin 2) is discharged when the IC transitions from skip to PWM mode. This forces the IC to remain in PWM mode for a fixed period of time. The added delay prevents unwanted switching between PWM and skip mode. The capacitor is charged with a 10uA current source on pin 2. The threshold on pin 2 is 2.5V. The delay for a typical 1nF capacitor is:
t delay =
CPWM x Vthreshold 1nF x 2.5V = = 250s Isource 10A
where: CPWM = capacitor connected to pin 2 Current Limit The current-limit circuit operates during PWM mode. The output current is detected by the voltage drop across the external current-sense resistor (RCS in Figure 2.). The cur-
The current-sense pins CSH (pin 8) and VOUT (pin 9) are noise sensitive due to the low signal level and high input impedance. The PCB traces should be short and routed close to each other. A small (1nF to 0.1F) capacitor across the pins will attenuate high frequency switching noise. When the peak inductor current exceeds the current-limit threshold, the current-limit comparator, in Figure 2, turns off the high-side MOSFET for the remainder of the cycle. The output voltage drops as additional load current is pulled from the converter. When the output voltage reaches approximately 0.95V, the circuit enters frequency-foldback mode and the oscillator frequency will drop to 60kHz while maintaining the peak inductor current equal to the nominal 100mV across the external current-sense resistor. This limits the maximum output power delivered to the load under a short circuit condition. Reference, Enable and, UVLO Circuits The output drivers are enabled when the following conditions are satisfied: * The VDD voltage (pin 11) is greater than its undervoltage threshold (typically 4.2V). * The voltage on the enable pin is greater than the enable UVLO threshold (typically 2.5V) The internal bias circuit generates a 1.245V bandgap reference voltage for the voltage error amplifier and a 5V VDD voltage for the gate drive circuit. The reference voltage in the fixed-output-voltage versions of the MIC2182 is buffered and brought to pin 7. The VREF pin should be bypassed to GND (pin 4) with a 0.1F capacitor. The adjustable version of the MIC2182 uses pin 7 for output voltage sensing. A decoupling capacitor on pin 7 is not used in the adjustable output voltage version.
Inductor Current
ILIM(skip) 0A
35mV THRESHOLD ACROSS RCS.
Figure 7a. Maximum Skip-Mode-Load Inductor Current
IMIN(PWM) Inductor Current 0A
12mV THRESHOLD OF AVERAGE VOLTAGE ACROSS RCS.
Figure 7b. Minimum PWM-Mode-Load Inductor Current for PWM Operation June 2000 13 MIC2182
MIC2182
The enable pin (pin 6) has two threshold levels, allowing the MIC2182 to shut down in a low current mode, or turn off output switching in UVLO mode. An enable pin voltage lower than the shutdown threshold turns off all the internal circuitry and reduces the input current to typically 0.1A. If the enable pin voltage is between the shutdown and UVLO thresholds, the internal bias, VDD, and reference voltages are turned on. The soft-start pin is forced low by an internal discharge MOSFET. The output drivers are inhibited from switching and remain in a low state. Raising the enable voltage above the UVLO threshold of 2.5V allows the softstart capacitor to charge and enables the output drivers. Either of two UVLO conditions will pull the soft-start capacitor low. * When the VDD drops below 4.1V * When the enable pin drops below the 2.5V threshold MOSFET Gate Drive The MIC2182 high-side drive circuit is designed to switch an N-channel MOSFET. Referring to the block diagram in Figure 2, a bootstrap circuit, consisting of D2 and CBST, supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the VSW pin (pin 15) is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the MOSFET turns on, the voltage on the VSW pin increases to approximately VIN. Diode D2 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. When the low-side switch is turned back on, CBST is recharged through D2. The drive voltage is derived from the internal 5V VDD bias supply. The nominal low-side gate drive voltage is 5V and the nominal high-side gate drive voltage is approximately 4.5V due the voltage drop across D2. A fixed 80ns delay between the high- and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs.
Micrel
Oscillator and Sync The internal oscillator is free running and requires no external components. The nominal oscillator frequency is 300kHz. If the output voltage is below approximately 0.95V, the oscillator operates in a frequency-foldback mode and the switching frequency is reduced to 60kHz. The SYNC input (pin 5) allows the MIC2182 to synchronize with an external clock signal. The rising edge of the sync signal generates a reset signal in the oscillator, which turns off the low-side gate drive output. The high-side drive then turns on, restarting the switching cycle. The sync signal is inhibited when the controller operates in skip mode or during frequency foldback. The sync signal frequency must be greater than the maximum specified free running frequency of the MIC2182. If the synchronizing frequency is lower, double pulsing of the gate drive outputs will occur. When not used, the sync pin must be connected to ground. Figure 8 shows the timing between the external sync signal (trace 2), the low-side drive (trace 1) and the high-side drive (trace R1). There is a delay of approximately 250ns between the rising edge of the external sync signal and turnoff of the low-side MOSFET gate drive. Some concerns of operating at higher frequencies are: * Higher power dissipation in the internal VDD regulator. This occurs because the MOSFET gates require charge to turn on the device. The average current required by the MOSFET gate increases with switching frequency. This increases the power dissipated by the internal VDD regulator. Figure 10 shows the total gate charge which can be driven by the MIC2182 over the input voltage range, for different values of switching frequency. The total gate charge includes both the high- and low-side MOSFETs. The larger SOP package is capable of dissipating more power than the SSOP package and can drive larger MOSFETs with higher gate drive requirements.
LOW-SIDE HIGH-SIDE DRIVE DRIVE
SYNC SIGNAL
TIME
VSS
VOUT
TIME
Figure 8. Sync Waveforms MIC2182 14
Figure 9. Startup Waveforms June 2000
MIC2182
* Reduced maximum duty cycle due to switching transition times and constant delay times in the controller. As the switching frequency increased, the switching period decreases. The switching transition times and constant delays in the MIC2182 start to become noticeable. The effect is to reduce the maximum duty cycle of the controller. This will cause the minimum input to output differential voltage (dropout voltage) to increase.
100 SOP
Micrel
The soft-start voltage is applied directly to the PWM comparator. A 5uA internal current source is used to charge up the soft-start capacitor. The capacitor is discharged when either the enable voltage drops below the UVLO threshold (2.5V) or the VDD voltage drops below the UVLO level (4.1V). The part switches at a minimum duty cycle when the soft-start pin voltage is less than 0.4V. This maintains a charge on the bootstrap capacitor and insures high-side gate drive voltage. As the soft-start voltage rises above 0.4V, the duty cycle increases from the minimum duty cycle to the operating duty cycle. The oscillator runs at the foldback frequency of 60kHz until the output voltage rises above 0.95V. Above 0.95V, the switching frequency increases to 300kHz (or the sync'd frequency), causing the output voltage to rise a greater rate. The rise time of the output is dependent on the soft-start capacitor, output capacitance, output voltage, and load current. The oscilloscope photo in Figure 9 show the output voltage and the soft-start pin voltage at startup. Minimum Pulse Width The MIC2182 has a specified minimum pulse width. This minimum pulse width places a lower limit on the minimum duty cycle of the buck converter. When the MIC2182 is operating in forced PWM mode (pin 2 low) and when the output current is very low or zero, there is a limit on the ratio of VOUT/VIN. If this limit is exceeded, the output voltage will rise above the regulated voltage level. A minimum load is required to prevent the output from rising up. This will not occur for output voltages greater than 3V. Figure 11 should be used as a guide when the MIC2182 is forced into PWM-only mode. The actual maximum input voltage will depend on the exact external components used (MOSFETs, inductors, etc.).
35
GATE CHARGE (nC)
80 60 40 400kHz 20 0 500kHz 300kHz
0
4
8 12 16 20 24 28 32 SUPPLY VOLTAGE (V)
Figure 10a. SOP Gate Charge vs. Input Voltage
100 SSOP
GATE CHARGE (nC)
80 60 300kHz 40 20 0 400kHz 500kHz 0 4 8 12 16 20 24 28 32 SUPPLY VOLTAGE (V)
Figure 10b. SSOP Gate Charge vs. Input Voltage It is recommended that the user limits the maximum synchronized frequency to 600kHz. If a higher synchronized frequency is required, it may be possible and will be design dependent. Please consult Micrel applications for assistance. Soft Start Soft start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitance is charged up. A slower output rise time will draw a lower input surge current. Soft start may also be used for power supply sequencing.
INPUT VOLTAGE (V)
30 25 20 15 10
0
1 2 3 4 5 OUTPUT VOLTAGE (V)
6
Figure 11. Max. Input Voltage in Forced-PWM Mode This restriction does not occur when the MIC2182 is set to automatic mode (pin 2 connected to a capacitor) since the converter operates in skip mode at low output current.
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MIC2182
MIC2182
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output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below: Pinductor Cu = Iinductor(rms)2 x R winding The resistance of the copper wire, Rwinding, increases with temperature. The value of the winding resistance used should be at the operating temperature. R winding(hot) = R winding(20C) x 1 + 0.0042 x (Thot - T20C )
Applications Information
The following applications information includes component selection and design guidelines. See Figure 14 and Tables 1 through 5 for predesigned circuits. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak to peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak to peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak to peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below.
(
)
L=
VOUT x (VIN(max) - VOUT ) VIN(max) x fS x 0.2 x IOUT(max)
where: fS = switching frequency 0.2 = ratio of ac ripple current to dc output current VIN(max) = maximum input voltage The peak-to-peak inductor current (ac ripple current) is:
IPP =
VOUT x (VIN(max) - VOUT ) VIN(max) x fS x L
where: THOT = temperature of the wire under operating load T20C = ambient temperature Rwinding(20C) is room temperature winding resistance (usually specified by the manufacturer) Current-Sense Resistor Selection Low inductance power resistors, such as metal film resistors should be used. Most resistor manufacturers make low inductance resistors with low temperature coefficients, designed specifically for current-sense applications. Both resistance and power dissipation must be calculated before the resistor is selected. The value of RSENSE is chosen based on the maximum output current and the maximum threshold level. The power dissipated is based on the maximum peak output current at the minimum overcurrent threshold limit.
RSENSE =
75mV IOUT(max)
The peak inductor current is equal to the average output current plus one half of the peak to peak inductor ripple current.
IPK = IOUT(max) + 0.5 x IPP
The maximum overcurrent threshold is:
Iovercurrent(max) = 135mV RCS
The RMS inductor current is used to calculate the I2*R losses in the inductor. 1 IPP Iinductor(rms) = IOUT(max) x 1 + 3 IOUT(max)
2
The maximum power dissipated in the sense resistor is:
PD(R
SENSE )
= Iovercurrent(max)2 x RCS
Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2182 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower
MOSFET Selection External N-channel logic-level power MOSFETs must be used for the high- and low-side switches. The MOSFET gateto-source drive voltage of the MIC2182 is regulated by an internal 5V VDD regulator. Logic-level MOSFETs, whose operation is specified at VGS = 4.5V must be used. It is important to note the on-resistance of a MOSFET increases with increasing temperature. A 75C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25C. This change in resistance must be accounted for when calculating MOSFET power dissipation. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2182 gate drive circuit. At 300kHz switching frequency and above, the gate 16 June 2000
MIC2182
MIC2182
charge can be a significant source of power dissipation in the MIC2182. At low output load this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is:
IG[high-side](avg) = QG x fS
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tT = CISS x VGS + COSS x VIN IG
where: IG[high-side](avg) = average high-side MOSFET gate current QG = total gate charge for the high-side MOSFET taken from manufacturer's data sheet with VGS = 5V. The low-side MOSFET is turned on and off at VDS = 0 because the freewheeling diode is conducting during this time. The switching losses for the low-side MOSFET is usually negligable. Also, the gate drive current for the lowside MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge. For the low-side MOSFET:
IG[low-side](avg) = CISS x VGS x fS
where: CISS and COSS are measured at VDS = 0. IG = gate drive current (1A for the MIC2182) The total high-side MOSFET switching loss is:
PAC = (VIN +VD ) x IPK x t T x fS
where: tT = switching transition time (typically 20ns to 50ns) VD = freewheeling diode drop, typically 0.5V. fS it the switching frequency, nominally 300kHz The low-side MOSFET switching losses are negligible and can be ignored for these calculations. RMS Current and MOSFET Power Dissipation Calculation Under normal operation, the high-side MOSFET's RMS current is greatest when VIN is low (maximum duty cycle). The low-side MOSFET's RMS current is greatest when VIN is high (minimum duty cycle). However, the maximum stress the MOSFETs see occurs during short circuit conditions, where the output current is equal to Iovercurrent(max). (See the Sense Resistor section). The calculations below are for normal operation. To calculate the stress under short circuit conditions, substitute Iovercurrent(max) for IOUT(max). Use the formula below to calculate D under short circuit conditions. Dshort circuit = 0.063 - 1.8 x 10 -3 x VIN The RMS value of the high-side switch current is:
ISW(highside)(rms) = I 2 D x IOUT(max)2 + PP 12
Since the current from the gate drive comes from the input voltage, the power dissipated in the MIC2182 due to gate drive is: Pgate drive = VIN IG[high-side](avg) + IG[low-side](avg)
(
)
A convenient figure of merit for switching MOSFETs is the onresistance times the total gate charge (RDS(on) x QG). Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2182. Power dissipation in the MIC2182 package limits the maximum gate drive current. Refer to Figure 10 for the MIC2182 gate drive limits. Parameters that are important to MOSFET switch selection are: * Voltage rating * On-resistance * Total gate charge The voltage rating of the MOSFETs are essentially equal to the input voltage. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitics. The power dissipated in the switching transistor is the sum of the conduction losses during the on-time (Pconduction) and the switching losses that occur during the period of time when the MOSFETs turn on and off (PAC).
ISW(low side)(rms) =
(1- D) IOUT(max)2 +
IPP2 12
where: D = duty cycle of the converter
D= x VIN VOUT
PSW = Pconduction + PAC where:
Pconduction = ISW(rms)2 x RSW PAC = PAC(off) + PAC(on)
RSW = on-resistance of the MOSFET switch. Making the assumption the turn-on and turnoff transition times are equal, the transition time can be approximated by: June 2000 17
= efficiency of the converter. Converter efficiency depends on component parameters, which have not yet been selected. For design purposes, an efficiency of 90% can be used for VIN less than 10V and 85% can be used for VIN greater than 10V. The efficiency can be more accurately calculated once the design is complete. If the assumed efficiency is grossly inaccurate, a second iteration through the design procedure can be made. For the high-side switch, the maximum dc power dissipation is: Pswitch1(dc) = RDS(on)1 x ISW1(rms)2
MIC2182
MIC2182
For the low-side switch (N-channel MOSFET), the dc power dissipation is: Pswitch2(dc) = RDS(on)2 x ISW 2(rms)2 Since the ac switching losses for the low side MOSFET is near zero, the total power dissipation is:
Plow-side MOSFET(max) = Pswitch2(dc)
Micrel
circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending on the circuit components and operating conditions, an external Schottky diode will give a 1/2% to 1% improvement in efficiency. Figure 12 illustrates the difference in noise on the VSW pin with and without a Schottky diode. Output Capacitor Selection The output capacitor values are usually determined by the capacitors ESR (equivalent series resistance). Voltage rating and RMS current capability are two other important factors in selecting the output capacitor. Recommended capacitors are tantalum, low-ESR aluminum electrolytics, and OS-CON. The output capacitor's ESR is usually the main cause of output ripple. The maximum value of ESR is calculated by:
RESR VOUT IPP
The total power dissipation for the high side MOSFET is:
PhighsideMOSFET(max) = PSWITCH 1(dc) + PAC
External Schottky Diode An external freewheeling diode is used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 80ns The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current.
ID(avg) = IOUT x 2 x 80ns x fS
The reverse voltage requirement of the diode is:
Vdiode(rrm) = VIN
The power dissipated by the Schottky diode is:
Pdiode = ID(avg) x VF
where: VF = forward voltage at the peak diode current The external Schottky diode, D2, is not necessary for circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the
WITHOUT WITH FREEWHEELING DIODE FREEWHEELING DIODE
where: VOUT = peak to peak output voltage ripple IPP = peak to peak inductor ripple current The total output ripple is a combination of the ESR and the output capacitance. The total ripple is calculated below: VOUT = IPP x (1- D) C + IPP x RESR OUT x fS
2
(
)2
where: D = duty cycle COUT = output capacitance value fS = switching frequency The voltage rating of capacitor should be twice the output voltage for a tantalum and 20% greater for an aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated below: IC (rms) = IPP 12
(rms)2 x RESR(C
OUT
The power dissipated in the output capacitor is:
PDISS(C
OUT )
= IC
OUT
OUT )
TIME
Figure 12. Switch Output Noise With and Without Shottky Diode MIC2182 18
Input Capacitor Selection The input capacitor should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. Tantalum input capacitor voltage rating should be at least 2 times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. June 2000
MIC2182
The input voltage ripple will primarily depend on the input capacitors ESR. The peak input current is equal to the peak inductor current, so:
Micrel
* Supply current to the MIC2182 * MOSFET gate-charge power (included in the IC supply current) * Core losses in the output inductor To maximize efficiency at light loads: * Use a low gate-charge MOSFET or use the smallest MOSFET, which is still adequate for maximum output current. * Allow the MIC2182 to run in skip mode at lower currents. * Use a ferrite material for the inductor core, which has less core loss than an MPP or iron power core. Under heavy output loads the significant contributors to power loss are (in approximate order of magnitude): * Resistive on-time losses in the MOSFETs * Switching transition losses in the MOSFETs * Inductor resistive losses * Current-sense resistor losses * Input capacitor resistive losses (due to the capacitors ESR) To minimize power loss under heavy loads: * Use logic-level, low on-resistance MOSFETs. Multiplying the gate charge by the on-resistance gives a Figure of merit, providing a good balance between low and high load efficiency. * Slow transition times and oscillations on the voltage and current waveforms dissipate more power during turn-on and turnoff of the MOSFETs. A clean layout will minimize parasitic inductance and capacitance in the gate drive and high current paths. This will allow the fastest transition times and waveforms without oscillations. Low gate-charge MOSFETs will transition faster than those with higher gate-charge requirements. * For the same size inductor, a lower value will have fewer turns and therefore, lower winding resistance. However, using too small of a value will require more output capacitors to filter the output ripple, which will force a smaller bandwidth, slower transient response and possible instability under certain conditions. * Lowering the current-sense resistor value will decrease the power dissipated in the resistor. However, it will also increase the overcurrent limit and will require larger MOSFETs and inductor components. * Use low-ESR input capacitors to minimize the power dissipated in the capacitors ESR. Decoupling Capacitor Selection The 4.7F decoupling capacitor is used to minimize noise on the VDD pin. The placement of this capacitor is critical to the proper operation of the IC. It must be placed right next to the
VIN = Iinductor(peak) x RESR(C
IN )
The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak to peak inductor ripple current is low:
IC (rms) IOUT(max) x
IN
D x (1- D)
The power dissipated in the input capacitor is:
PDISS(C ) = IC (rms)2 x RESR(C ) IN IN IN
Voltage Setting Components The MIC2182-3.3 and MIC2182-5.0 ICs contain internal voltage dividers that set the output voltage. The MIC2182 adjustable version requires two resistors to set the output voltage as shown in Figure 13.
R1 Error Amp FB
7
R2
VREF 1.245V MIC2182 [adj.]
Figure 13. Voltage-Divider Configuration The output voltage is determined by the equation: R1 VO = VREF x 1 + R2 Where: VREF for the MIC2182 is typically 1.245V. A typical value of R1 can be between 3k and 10k. If R1 is too large it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value it will decrease the efficiency of the power supply, especially at low output loads. Once R1 is selected, R2 can be calculated using:
R2 = VREF x R1 VO - VREF
Voltage Divider Power Dissipation
The reference voltage and R2 set the current through the voltage divider.
V Idivider = REF R2 The power dissipated by the divider resistors is:
Pdivider = (R1+ R2) x Idivider 2
Efficiency Calculation and Considerations Efficiency is the ratio of output power to input power. The difference is dissipated as heat in the buck converter. Under light output load, the significant contributors are:
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MIC2182
MIC2182
pins and routed with a wide trace. The capacitor should be a good quality tantalum. An additional 1F ceramic capacitor may be necessary when driving large MOSFETs with high gate capacitance. Incorrect placement of the VDD decoupling capacitor will cause jitter or oscillations in the switching waveform and large variations in the overcurrent limit. A 0.1F ceramic capacitor is required to decouple the VIN. The capacitor should be placed near the IC and connected directly to between pin 10 (Vcc) and pin 12 (PGND). PCB Layout and Checklist PCB layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the circuit. * Signal and power grounds should be kept separate and connected at only one location. Large currents or high di/dt signals that occur when the MOSFETs turn on and off must be kept away from the small signal connections. * The connection between the current-sense resistor and the MIC2182 current-sense inputs (pin 8 and 9) should have separate traces, routed from the terminals directly to the IC pins. The traces should be routed as closely as possible to each other and their length should be minimized. Avoid running the traces under the inductor and other switching components. A 1nF to 0.1F capacitor placed between pins 8 and 9 will help attenuate switching noise on the current sense traces. This capacitor should be placed close to pins 8 and 9.
Micrel
* When the high-side MOSFET is switched on, the critical flow of current is from the input capacitor through the MOSFET, inductor, sense resistor, output capacitor, and back to the input capacitor. These paths must be made with short, wide pieces of trace. It is good practice to locate the ground terminals of the input and output capacitors close to each. * When the low-side MOSFET is switched on, current flows through the inductor, sense resistor, output capacitor, and MOSFET. The source of the low-side MOSFET should be located close to the output capacitor. * The freewheeling diode, D1 in Figure 2, conducts current during the dead time, when both MOSFETs are off. The anode of the diode should be located close to the output capacitor ground terminal and the cathode should be located close to the input side of the inductor. * The 4.7F capacitor, which connects to the VDD terminal (pin 11) must be located right at the IC. The VDD terminal is very noise sensitive and placement of this capacitor is very critical. Connections must be made with wide trace. The capacitor may be located on the bottom layer of the board and connected to the IC with multiple vias. * The VIN bypass capacitor should be located close to the IC and connected between pins 10 and 12. Connections should be made with a ground and power plane or with short, wide trace.
MIC2182
20
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MIC2182
Predesigned Circuits A single schematic diagram, shown in Figure 14, can be used to build power supplies ranging from 3A to 10A at the common output voltages of 1.8V, 2.5V, 3.3V, and 5V. Components that vary, depending upon output current and voltage, are listed in the accompanying Tables 3 through 6.
Micrel
Power supplies larger than 10A can also be constructed using the MIC2182 using larger power-handling components. The "Power Supply Operating Characteristics" graphs following the component and vendor tables provide useful information about the actual performance of some of these circuits.
VIN R7 100k C5 0.1F
MIC2182 VIN VDD BST EN/UVLO HSD PWM C4 1nF SS COMP C3 0.1F C2 2.2nF R1 2k SYNC VSW LSD PGND CSH
D2 SD103BWS C6 0.1F C9 4.7F 16V
C11 (table)
Q2 (table) Q1 (table)
L1 (table) D1 (table)
R2 (table) C7 (table)
VOUT C12 0.1F 50V GND
C13, 1nF VOUT VREF SGND C1 0.1F 50V
GND
Figure 14. Basic Circuit Diagram for Use with Tables 3 through 6
Specification Switching frequency ripple Maximum ambient temperature Short-circuit capability Switching frequency
Limit 1% of output voltage 85C Continuous 300kHz
Table 1. Specifications for Figure 14 and Tables 3 through 6
Manufacturer AVX Central Semiconductor Coiltronics IRC IR Micrel Vishay/Lite On (diodes) Vishay/Siliconix (MOSFETs) Vishay/Dale (inductors and resistors) Sumida
Telephone Number (USA) (803) 946-0690 (516) 435-1110 (561) 241-7876 (704) 264-8861 (310) 322-3331 (408) 944-0800 (805) 446-4800 (800) 554-5665 (800) 487-9437 (847) 956-0666
Web Address www.avxcorp.com www.centralsemi.com www.coiltronics.com
www.irf.com www.micrel.com www.vishay-liteon.com www.siliconix.com www.vishaytechno.com www.japanlink.com/sumida
Table 2. Component Suppliers
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MIC2182
MIC2182
Reference C7 3A (6.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH125-100, Sumida Inductor, 10H 4A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .025 1%, Vishay, 0.025, 1%, 0.5W, current sense resistor MIC2182-5.0BSM or MIC2182-5.0BM 4A (6.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 3 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100, Sumida Inductor, 10H 5A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .020 1%, Vishay, 0.02, 1%, 0.5W, current sense resistor MIC2182-5.0BSM or MIC2182-5.0BM 5A (6.5V-30V) Part No. / Description qty: 2 TPSV227M010R0060 AVX, 220F 10V, 0.06 ESR, output filter capacitor qty: 4 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100 Sumida, 10H 5A, output inductor qty: 1 Si4884, Siliconix, low-side MOSFET qty: 1 Si4884, Siliconix, high-side MOSFET qty: 1 WSL-2512 .015 1%, Vishay, 0.015, 1%, 1W, current sense resistor MIC2182-5.0BSM or MIC2182-5.0BM
Micrel
10A (6.5V-10V) Part No. / Description qty: 2 TPSV337M010R0060 AVX, 330F 10V, 0.06 ESR, output filter capacitor qty: 4 TPSV107M020R0085 AVX, 100F 20V, 0.06 ESR, input filter capacitor qty: 1 B330, Vishay, freewheeling diode qty: 1 UP4B-3R3, Coiltronics, 3.3H 11A, output inductor qty: 2 Si4884, Siliconix low-side MOSFET qty: 2 Si4884, Siliconix, high-side MOSFET qty: 2 WSL-2512 .015 1% , Vishay, 0.015, 1%, 1W, current sense resistor MIC2182-5.0BM
C11
D1 L1
Q1 Q2 R2
U1
Table 3. Components for 5V Output
Reference C7
3A (4.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH125-100, Sumida Inductor, 10H 4A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .025 1%, Vishay, 0.025, 1%, 0.5W, current sense resistor MIC2182-3.3BSM or MIC2182-3.3BM
4A (4.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100, Sumida Inductor, 10H 5A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .020 1%, Vishay, 0.02, 1%, 0.5W, current sense resistor MIC2182-3.3BM or MIC2182-3.3BSM
5A (4.5V-30V) Part No. / Description qty: 2 TPSV227M010R0060 AVX, 220F 10V, 0.06 ESR, output filter capacitor qty: 3 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100 Sumida, 10H 5A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4884, Siliconix, high-side MOSFET qty: 1 WSL-2512 .015 1%, Vishay, 0.015, 1%, 1W, current sense resistor MIC2182-3.3BM or MIC2182-3.3BSM
10A (4.5V-5.5V) Part No. / Description qty: 2 TPSV477M006R0055 AVX, 470F 6.3V, 0.055 ESR, output filter capacitor qty: 3 TPSV227M016R0075 AVX, 220F 16V, 0.075 ESR, filter capacitor qty: 1 B330, Vishay, freewheeling diode qty: 1 UP4B-3R3, Coiltronics, 3.3H 11A, output inductor qty: 2 Si4884, Siliconix, low-side MOSFET qty: 2 Si4884, Siliconix, high-side MOSFET qty: 2 WSL-2512 .015 1% , Vishay, 0.015, 1%, 1W, current sense resistor MIC2182-3.3BM
C11
D1 L1
Q1 Q2 R2
U1
Table 4. Components for 3.3V Output MIC2182 22 June 2000
MIC2182
Reference C7 3A (4.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH125-100, Sumida Inductor, 10H 4A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .025 1%, Vishay, 0.025, 1%, 0.5W, current sense resistor MIC2182BSM or MIC2182BM 4A (4.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100, Sumida Inductor, 10H 5A, output inductor qty: 1 Si4884, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .020 1%, Vishay, 0.02, 1%, 0.5W, current sense resistor MIC2182BSM or MIC2182BM 5A (4.5V-30V) Part No. / Description qty: 2 TPSV227M010R0060 AVX, 220F 10V, 0.06 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100 Sumida, 10H 5A, output inductor qty: 1 Si4884, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2512 .015 1%, Vishay, 0.015, 1%, 1W, current sense resistor MIC2182BSM or MIC2182BM
Micrel
10A (4.5V-5.5V) Part No. / Description qty: 2 TPSV447M006R0055 AVX, 470F 6.3V, 0.06 ESR, output filter capacitor qty: 3 TPSV227M016R0075 AVX, 220F 16V, 0.06 ESR, input filter capacitor qty: 1 B330, Vishay, freewheeling diode qty: 1 UP4B-3R3, Coiltronics, 3.3H 11A, output inductor qty: 2 Si4884, Siliconix low-side MOSFET qty: 2 Si4884, Siliconix, high-side MOSFET qty: 1 WSL-2512 .015 1% , Vishay, 0.015, 1%, 1W, current sense resistor MIC2182BM
C11
D1 L1
Q1 Q2 R2
U1
Table 5. Components for 2.5V Output
Reference C7
3A (4.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH125-100, Sumida Inductor, 10H 4A, output inductor qty: 1 Si4800, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .025 1%, Vishay, 0.025, 1%, 0.5W, current sense resistor MIC2182BSM or MIC2182BM
4A (4.5V-30V) Part No. / Description qty: 2 TPSE227M010R0100 AVX, 220F 10V, 0.1 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100, Sumida Inductor, 10H 5A, output inductor qty: 1 Si4884, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2010 .020 1%, Vishay, 0.02, 1%, 0.5W, current sense resistor MIC2182BSM or MIC2182BM
5A (4.5V-8V) Part No. / Description qty: 2 TPSV227M010R0060 AVX, 220F 10V, 0.06 ESR, output filter capacitor qty: 2 TPSE226M035R0300 AVX, 22F 35V, 0.3 ESR, input filter capacitor qty: 1 B140, Vishay, freewheeling diode qty: 1 CDRH127-100 Sumida, 10H 5A, output inductor qty: 1 Si4884, Siliconix, low-side MOSFET qty: 1 Si4800, Siliconix, high-side MOSFET qty: 1 WSL-2512 .015 1%, Vishay, 0.015, 1%, 1W, current sense resistor MIC2182BSM or MIC2182BM
10A (4.5V-5.5V) Part No. / Description qty: 2 TPSV447M006R0055 AVX, 470F 6.3V, 0.06 ESR, output filter capacitor qty: 2 TPSV227M016R0075 AVX, 220F 16V, 0.06 ESR, input filter capacitor qty: 1 B330, Vishay, freewheeling diode qty: 1 UP4B-3R3, Coiltronics, 3.3H 11A, output inductor qty: 2 Si4884, Siliconix low-side MOSFET qty: 2 Si4884, Siliconix, high-side MOSFET qty: 2 WSL-2512 .015 1% , Vishay, 0.015, 1%, 1W, current sense resistor MIC2182BM
C11
D1 L1
Q1 Q2 R2
U1
Table 6. Components for 1.8V Output June 2000 23 MIC2182
MIC2182
Power Supply Operating Characteristics
Effect of Soft-Start Capacitor (CSS) Value On Output Voltage Waveforms During Turn-On (10A Power Supply Configuration)
Micrel
Effect of Soft-Start Capacitor (CSS) Value On Output Voltage Waveforms During Turn-On (4A Power Supply Configuration)
Normal (300kHz Switching Frequency) and Output Short-Circuit (60kHz) Conditions Switch Node (Pin 15) Waveforms
Converter Waveforms
VSW PIN 15
SWITCH-NODE VOLTAGE
VSW+HSD
PIN 16
HIGH-SIDE DRIVE VOLTAGE REFERENCED TO GROUND
VIN = 7V L1 = 3.3H VOUT = 3.3V IOUT = 10A
VGS HIGH-SIDE MOSFET
HIGH-SIDE MOSFET GATE-TO-SOURCE VOLTAGE LOW-SIDE MOSFET GATE-TO-SOURCE VOLTAGE
QTY: 2 Si4884 HIGH-SIDE MOSFETS QTY: 2 Si4884 LOW-SIDE MOSFETS
IL1 (2A/div)
VGS LOW-SIDE MOSFET
INDUCTOR CURRENT
10Amps
Typical Skip-Mode Waveforms
Typical PWM-Mode Waveforms
VOUT
VSW Pin 15
MIC2182
IL1 (0.5A/div)
24
IL1 (0.5A/div)
VSW Pin 15
VOUT
June 2000
MIC2182
Micrel
Load Transient Response and Bode Plot (4A Power Supply Configuration)
Load Transient Response and Bode Plot (10A Power Supply Configuration)
VOUT
VIN = 12V VOUT = 3.3V L1 = 10H R2 = 20m
IOUT 5A/div
VOUT
VIN = 6V VOUT = 3.3V L1 = 3.3H R2 = 7.5m
IOUT 2A/div
Bode Plot (4A Power Supply Configuration)
100 80 60
GAIN (dB)
210 180
5V Efficiency (4A Power Supply Configuration)
100 Skip 80 EFFICIENCY (%)
PHASE ()
Bode Plot (10A Power Supply Configuration)
100 80 60 GAIN 210 180 150 120 90 60 PHASE
100x100 1x103 100x103 10x103 PHASE ()
PWM
GAIN (dB)
GAIN PHASE
150 120 90 60 30
40 20 0 -20
100x100 1x103
60 40 VIN = 5V R2 = 15m L1 = 10H 20 1 high-side MOSFET: Si4800 1 low-side MOSFET: Si4800 0 0.01 0.1 1 4 OUTPUT CURRENT (A)
40 20 0 -20
10x100
30
300x103
100x103
FREQUENCY (Hz)
300x103
10x100
10x103
-40
0
-40
0
FREQUENCY (Hz)
12V Efficiency (4A Power Supply Configuration)
100 80 Skip 60 40 VIN = 12V R2 = 15m L1 = 10H 20 1 high-side MOSFET: Si4800 1 low-side MOSFET: Si4800 0 0.01 0.1 1 4 OUTPUT CURRENT (A) PWM EFFICIENCY (%)
24V Efficiency (4A Power Supply Configuration)
100 80 Skip 60 40 VIN = 24V R2 = 15m L1 = 10H 20 1 high-side MOSFET: Si4800 1 low-side MOSFET: Si4800 0 0.01 0.1 1 4 OUTPUT CURRENT (A)
Efficiency (10A Power Supply Configuration)
100 Skip 80 EFFICIENCY (%)
EFFICIENCY (%)
PWM
60 40 20
PWM
R2 = 7.5m L1 = 3.3H 2 high-side MOSFETs: Si4884 2 low-side MOSFETs: Si4884
0 0.01
0.1 1 OUTPUT CURRENT (A)
10
June 2000
25
MIC2182
MIC2182
Micrel
Package Information
PIN 1
0.157 (3.99) 0.150 (3.81)
DIMENSIONS: INCHES (MM)
0.020 (0.51) REF 0.050 (1.27) BSC
0.020 (0.51) 0.013 (0.33) 0.0098 (0.249) 0.0040 (0.102)
45 0-8 0.050 (1.27) 0.016 (0.40) 0.244 (6.20) 0.228 (5.79)
0.0648 (1.646) 0.0434 (1.102)
0.394 (10.00) 0.386 (9.80)
SEATING PLANE
16-pin SOP (M)
5.40 (0.213) 5.20 (0.205) 7.90 (0.311) 7.65 (0.301)
DIMENSIONS: MM (INCH)
0.875 (0.034) REF 6.33 (0.239) 6.07 (0.249) 2.00 (0.079) 1.73 (0.068)
10 4
0.22 (0.009) 0.13 (0.005)
0.38 (0.015) 0.25 (0.010)
0.21 (0.008) 0.05 (0.002) 0.65 (0.0260) COPLANARITY: 0.10 (0.004) MAX BSC
0 -8
1.25 (0.049) REF 0.95 (0.037) 0.55 (0.022)
16-Pin SSOP (SM)
MIC2182
26
June 2000
MIC2182
Micrel
June 2000
27
MIC2182
MIC2182
Micrel
MICREL INC. 1849 FORTUNE DRIVE
TEL
SAN JOSE, CA 95131
WEB
USA
+ 1 (408) 944-0800
FAX
+ 1 (408) 944-0970
http://www.micrel.com
This information is believed to be accurate and reliable, however no responsibility is assumed by Micrel for its use nor for any infringement of patents or other rights of third parties resulting from its use. No license is granted by implication or otherwise under any patent or patent right of Micrel Inc. (c) 2000 Micrel Incorporated
MIC2182
28
June 2000

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